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Extreme Conversion with Fast Transient Response

Wed, 04/23/2014 - 1:18pm
Vince Stueve, FAE & Jayant Rao, Micrel, Inc.

Flyback Controllers, Push-Pull and Dual Stage Conversion are traditional methods used when faced with the task of converting rails of greater than 48V down to 3.3V or less. This large conversion ratio (step-down) is extreme and requires tight control. What if a simplified, straightforward buck regulator were able to be used for this extreme conversion? How much simpler could the circuit design be?

Imagine how much real estate and cost could be saved if a simple "off the shelf" inductor could be used rather than a custom transformer like those used in Flyback or Push-Pull topologies? Then imagine that there is no need for compensation as the load impedance changes and all of this is done with a fast load transient response. Sounds too good to be true?

This scenario is actually made possible using DC/DC regulator technology that is new to the marketplace. This article will explain this technology, introduce a unique patented ripple injection method and provide simple design equations for ensuring proper operation.

Dual Operation Modes: Controllers that use Adaptive ON-Time Control are employed for fast load transient response. Their behavior can be explained in the following two operation modes:

  1. During steady-state operation the part behaves very similar to a traditional Voltage Mode PWM (Pulse Width Modulated) regulator. The integrated ON-Time Estimator, along with Voltage feedback into an error comparator, generates ON-Time periods for the High Side MOSFET. 

For a Buck regulator, FSW = VOUT/(VIN*TON). The On-Time estimator produces On-Time pulses that are inversely proportional to Input Voltage, VIN, and directly proportional to Output Voltage, VOUT, so as to keep the switching frequency, FSW, a constant. The resulting switching frequency varies no more than +10 percent over input line changes allowing for easy filtering of power supply noise. 

  1. At the time of large load transients, the output voltage can take a large dip and the second operation mode kicks in. The Adaptive On-Time architecture drives the High Side MOSFET with rapid bursts of energy during a load transient to assist the output Inductor/Capacitor (LC) stage in keeping up with load requirements. These rapid "Hyperactive" pulses are created by digitally modifying the OFF-Time of the High Side MOSFET work to recover the lost energy incurred by a load transient and keep the output voltage in regulation.

The SuperSwitcher II™ family of regulators implements new DC/DC regulator technology built around a controller that uses Adaptive ON-Time Control and employs Hyper Speed Control™ Architecture. This new family of integrated Synchronous Buck Regulators can operate with input voltages from 4.5V to 70V thus enabling extreme conversions with fast load transient response. Output voltages down to 0.8V and currents of 4A are now able to be realized in a simple single stage integrated Buck Regulator housed in a small 3mm x 4mm MLF package.

Figure 1 compares ON-Time of the High Side MOSFET in a traditional Voltage mode PWM controller to the Adaptive ON-Time SuperSwitcher II architecture. Traditional Voltage mode PWM controllers require more time to respond to a load transient because they run as fixed PWMs. In a traditional voltage mode PWM, the engineer can counteract the dip in output voltage by adding more capacitance to the output. The additional capacitance needs to have a low Equivalent Series Resistance (ESR) to keep up with the load transient. One way of minimizing ESR is to stack many Ceramic capacitors in parallel on the output stage of the regulator and then add additional capacitance around the board to ensure proper operation. This represents a fairly common practice. Additionally, the storage capability (inductance) of the inductor can be increased. Together these additions to a traditional PWM regulator will increase storage capability of the LC portion of the circuit thus attempting to keep output voltage steady when load transients occur.Figure 1.  SuperSwitcher II Load Transient Operation Versus Traditional Voltage Mode PWM.

Consider a single stage conversion of 48V to 1.2V; this requires a duty cycle of 1.2V/48V = 4 percent. A duty cycle of 4 percent means the top side MOSFET of the switching regulator is ON only 4 percent of the time while the bottom side MOSFET is ON 96 percent of the time. In such a conversion if a traditional PWM controller is used then a large amount of capacitance is required to keep up with the load transient because new additional charge for the load can only occur when the top side MOSFET is ON.

Using this new approach, the magnitude of the output LC can be reduced because the controller inserts energy into the LC at exactly the right time. Now, smaller groupings or even a single very low ESR Ceramic Capacitor along with reasonably sized output inductors can be used to keep up with load transients. The smaller scale of the output stage coupled with the integration of MOSFETs into the regulator allows for minimal real estate to be occupied by the POL regulator solution. All of this helps to save real estate on the PCB, simplifies layout, and reduces total system cost.

Stability: Cumbersome compensation that is commonplace in traditional voltage mode PWM’s has been addressed using a unique method of Ripple Injection at the feedback node. For stability of an Adaptive-On Time controller, the Feedback Comparator must have sufficient overdrive to operate and thus a ripple voltage (between 20mV to 100mV) is required on the FB node. It's also required that this feedback ripple voltage be in phase with the inductor current to prevent switching harmonics.

"Off the shelf" inductors can now be used as long as enough ripple injection exists for proper operation. Once ripple injection is properly placed on the feedback node, any additional capacitance added downstream of the regulator will not affect loop stability of the circuit. This means that additional output capacitance, in the form of tantalum, electrolytic or other decoupling capacitors which might be distributed around the PCB at layout time, will not affect the stability of the regulator. This simple yet highly effective means of stabilizing the regulator frees the power supply designer from the worry that digital or system designers might kludge extra capacitance downstream of the regulator thereby affecting its stability.

The ESR of the output Capacitor will dictate how the ripple is brought into the feedback loop. Three methods of injecting ripple onto the feedback node are possible and depend on output capacitor ESR:

  1. If the output capacitor (Figure 2) has a large ESR as found in most Tantalum and Electrolytic capacitors, then the resulting ripple present at the feedback node is the divided down output ripple and is defined by Equation 1. Feedback ripple should be between 20mV to 100mV range to maintain stability of the DC/DC regulator. With large ESR, this ripple is triangular in shape and it is in phase with the inductor current.Figure 2.  Enough Ripple at Feedback Node, COUT ESR is Large. 

Equation 1:

ΔVFB(pp) = (R11/(R1+R11)) * ESRCOUT * ΔIL(pp)

Where: ΔIL(pp) is the peak-to-peak value of the inductor ripple current.

ΔVFB(pp) is the Peak to Peak ripple voltage on the Feedback node.

 

  1. If the output capacitor has an ESR that is small but not quite as low as a Ceramic capacitor, then the divided down ripple at the feedback node might not be adequate (less than 20mV). In this case a feed-forward capacitor CFF with a value in the range of 1nF to 100nF can be used to AC couple the output ripple into the Feedback node. This could be the case when using Tantalum capacitors.


Equation 2:

ΔVFB(pp) = ESRCOUT * ΔIL(pp)

 

  1. When Ceramic output capacitors are used the ESR is so low that virtually no ripple is present on the output and hence feedback node. This is the preferred method of using the new DC/DC regulator, so the resulting output voltage has a very low ripple. The ripple injection method introduced in Figure 4 uses the switch node to produce an AC ripple voltage on the FB node. Components Rinj and Cinj are used to generate an injection current that capacitor CFF integrates to produce a triangular ramp on the Feedback node.

Figure 4. Ceramic Capacitor on Output, Hence COUT ESR is Very Low.

Let one now consider a design example for a ceramic output capacitor with a net 5mΩ ESR and develop an understanding of this Ripple Injection technique. Consider the 70V Power Module MIC28304 for a VIN=48 , a VOUT=3., an IOUT=3A and at a 600kHz Switching Frequency (i.e., the Frequency select pin FREQ is left open). The MIC28304 module includes a 4.7μH inductor. Setting the output capacitor at a recommended 47μF Ceramic capacitor, the output ripple, which is the vector sum due to ESR of the output cap and inductor ripple current, is:

∆Vout(pp) = ΔIL(pp) *ESR + ΔIL(pp) /(8*FSW*COUT) ≈ 10mV

This is not enough ripple for proper operation hence ripple injection is needed. Figure 5 shows the circuit for the MIC28304 with the ripple injection scheme. Calculations for injection components are as follows:

1. Pick Cinj = 100nF so it’s an AC short for current injection. The Rinj and CFF values are calculated to ensures a recommended 50mV of feedback ripple ΔVFB(pp).

2. Choosing R1=10k, R2 is calculated to be 3.16k for VOUT =3.3V (given VREF = 0.8V).

As no average DC current can flow through Cinj, average voltage for node A is the same as SW node. Average voltage for SW node = VIN*D = VOUT = Average voltage for node A

3. During the ON time, SW node is at VIN. Thus, current injected into FB during ON time is:

IRinj(TON) = (VIN-VOUT)/Rinj ………………….(Equation 3)

4. Similarly, during the OFF time, the SW node is at GND and thus IRinj(TOFF) is the current removed.

Figure 6 illustrates the waveforms for current injection during TON and removal during TOFF.

Figure 5: MIC28304 with the Ripple Injection Scheme.

Figure 7 shows the AC model for this injection scheme. VOUT is assumed AC GND as it has little ripple.

It is desirable to have the majority of injected current to go through CFF so as to get more of a triangular ripple on VFB, in phase with the inductor current, as shown in Figure 6. To ensure this, Z(CFF) should be one-tenth of R1|| R2.

5. Thus setting Z(CFF) = (R1||R2)/10 @ FSW = 600kHz

ð To provide enough margin, we pick CFF ≈ 2.2nF

6. Now ∆VFB(pp) = (IRinj(TON) *TON )/ CFF ………………….(Equation 4)

Inserting value of IRinj(TON) from Equation 3, and using D=TON/FSW

ð Rinj = VIN D (1- D)/ (Fsw*∆VFB(pp) *CFF)

Substituting CFF = 2.2nF and ∆VFB(pp) = 50mV, we pick Rinj ≈ 51.1k as the appropriate standard resistor value available.

Figure 6. Waveforms Showing Injection During On and Off Times.

Figure 7. AC Model for Ripple Injection Scheme.

Steady State measurements show a feedback ripple of about 62mV. This is because the ripple from the Output capacitor gets AC-coupled through the capacitor CFF  and adds to the ripple injection from the SW node. This can be seen in the slightly curved part of the Feedback ripple waveform. As seen below, in Figure 8, the feedback ripple is in phase with the switch node and hence the internal inductor current.

Figure 8: Measurements for Injected Ripple.A scope shot illustrating the performance of this new family of DC/DC regulators during a load transient is shown in Figure 9. As was shown in Figure 1 earlier, the implemented control scheme modifies the OFF time to provide rapid pulses of energy to charge up the output LC quickly as the load transient occurs.

Figure 9.  0.5A to 3A Load Step showing Switch Node in Hyper Speed Control VIN = 48V, VOUT = 3.3V.Operation up to 70V for DC/DC regulators is groundbreaking. Micrel's SuperSwitcher II excels in these Extreme Conversions ratio as a straight buck while providing excellent line and load regulation. The ability to survive up to 70V is a major advantage and opens up Automotive and Industrial applications where the regulator might have a VIN of 12V or 36V but must survive input load transients that approach the 70V range. Automotive applications refer to these large transients to the 12V battery rail as Load Dump. The architecture is also very suitable for LED Buck Drivers and non-isolated PoE applications that run off 48V typical.

The remarkable load transient response achieved by these regulators can also be found in 28V and 36V versions. The 70V, MIC28304 highlighted in this article is made for extreme conversions, or as indicated to survive in conditions where the input voltage may reach levels much higher than expected due to events such as load dump. The lower voltage range parts have their place and should not be overlooked as a way to save system cost. Internal MOSFETs allow for output currents spanning the range of 4A to 12A in a typical 3mm x 4mm package. 

More information, Datasheets and Evaluation boards for the SuperSwitcher II family of regulators can be found on the Micrel web site thereby allowing the design of simple, efficient and cost effective Point of Load regulator circuits.

Note: SuperSwitcher II, Any Capacitor and Hyper Speed Control are trademarks of Micrel, Inc.

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